View AD8203_368908.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

High Common-Mode Voltage, Single-Supply Difference Amplifier AD8203
FEATURES
High common-mode voltage range -6 V to +30 V at a 5 V supply voltage Operating temperature range: -40C to +125C Supply voltage range: 3.5 V to 12 V Low-pass filter (1-pole or 2-pole) Excellent ac and dc performance 1 mV voltage offset (8-lead SOIC) 1 ppm/C typical gain drift 80 dB CMRR minimum dc to 10 kHz
FUNCTIONAL BLOCK DIAGRAMS
NC
6
A1
3
A2
4
+VS
7
AD8203
100k G = x7 +IN 8 -IN 1 200k +IN A1 -IN 200k 10k
05013-001
G = x2 +IN A2 -IN 10k
5
OUT
APPLICATIONS
Transmission control Diesel injection control Engine management Adaptive suspension control Vehicle dynamics control
NC = NO CONNECT
2
GND
Figure 1. Functional Block Diagram
CLAMP DIODE
INDUCTIVE 5V LOAD OUTPUT
+IN +VS NC OUT
GENERAL DESCRIPTION
The AD8203 is a single-supply difference amplifier for amplifying and low-pass filtering small differential voltages in the presence of a large common-mode voltage (CMV). The input CMV range extends from -6 V to +30 V at a typical supply voltage of 5 V. The AD8203 is available in die and packaged form. The MSOP and SOIC packages are specified over a wide temperature range, from -40C to +125C, while the die is specified over a wider temperature range, from -40C to +150C, making the AD8203 well-suited for use in many automotive platforms. Automotive platforms demand precision components for better system control. The AD8203 provides excellent ac and dc performance keeping errors to a minimum in the user's system. Typical offset and gain drift in the SOIC package are 0.3 V/C and 1 ppm/C, respectively. Typical offset and gain drift in the MSOP package are 2 V/C and 1 ppm/C, respectively. The device also delivers a minimum CMRR of 80 dB from dc to 10 kHz. The AD8203 features an externally accessible 100 k resistor at the output of the Preamp A1, which can be used for low-pass filter applications and for establishing gains other than 14.
BATTERY 14V 4-TERM SHUNT
AD8203
-IN GND A1 A2
POWER DEVICE
COMMON
NC = NO CONNECT
Figure 2. High Line Current Sensor
POWER DEVICE
5V OUTPUT
+IN
+VS
NC
OUT
BATTERY
14V 4-TERM SHUNT
AD8203
-IN GND A1 A2
CLAMP DIODE
INDUCTIVE LOAD
05013-003
COMMON
NC = NO CONNECT
Figure 3. Low Line Current Sensor
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c) 2005 Analog Devices, Inc. All rights reserved.
05013-002
AD8203 TABLE OF CONTENTS
Features .............................................................................................. 1 Applications....................................................................................... 1 Functional Block Diagrams............................................................. 1 Specifications..................................................................................... 3 Single Supply ................................................................................. 3 Absolute Maximum Ratings............................................................ 4 ESD Caution.................................................................................. 4 Pin Configuration and Function Descriptions............................. 5 Typical Performance Characteristics ............................................. 6 Theory of Operation ...................................................................... 12 Applications..................................................................................... 14 Current Sensing .......................................................................... 14 Gain Adjustment ........................................................................ 14 Gain Trim .................................................................................... 15 Low-Pass Filtering...................................................................... 15 High Line Current Sensing with LPF and Gain Adjustment ........................................................................ 16 Driving Charge Redistribution ADCs..................................... 16 Outline Dimensions ....................................................................... 17 Ordering Guide .......................................................................... 17
REVISION HISTORY
10/05--Rev. A to Rev. B Added SOIC Package .........................................................Universal Replaced Figure 23 ........................................................................... 8 Added Figure 24 to Figure 29.......................................................... 9 Changes to Theory of Operation Section ................................... 12 Added Figure 41.............................................................................. 12 Updated Outline Dimensions ....................................................... 17 Changes to Ordering Guide .......................................................... 17 2/05--Rev. 0 to Rev. A Changes to Specifications Table...................................................... 3 Changes to Caption on Figure 6 and Figure 8 .............................. 6 Changes to Figure 12........................................................................ 7 Added Figure 14 to Figure 23.......................................................... 7 Changes to Figure 26 and Figure 27............................................. 10 Changes to Figure 29...................................................................... 11 Changes to Figure 32 and Figure 33............................................. 12 Changes to Ordering Guide .......................................................... 13 10/04--Revision 0: Initial Version
Rev. B | Page 2 of 20
AD8203 SPECIFICATIONS
SINGLE SUPPLY
TA = operating temperature range, VS = 5 V, unless otherwise noted. Table 1.
Parameter SYSTEM GAIN Initial Error vs. Temperature VOLTAGE OFFSET Input Offset (RTI) vs. Temperature INPUT Input Impedance Differential Common Mode CMV CMRR 1 Conditions AD8203 SOIC Min Typ Max 14 0.02 VOUT 4.8 V dc @ 25C -0.3 1 VCM = 0.15 V; 25C -40C to +125C -40C to +150C -1 -10 +0.3 20 +1 +10 -0.3 1 -2 -20 AD8203 MSOP Min Typ Max 14 +0.3 25 +2 +20 -0.3 1 -1 -10 -15 Min AD8203 Die Typ Max 14 +0.3 30 +1 +10 +15 Unit V/V % ppm/C mV V/C V/C
+0.3
+2
+0.3 +5
Continuous VCM = -6 V to +30 V f = dc f = 1 kHz f = 10 kHz 2
260 130 -6 82 82 80
320 160
380 190 +30
260 130 -6 82 82 80
320 160
380 190 +30
260 130 -6 82 82 80
320 160
380 190 +30
k k V dB dB dB
PREAMPLIFIER Gain Gain Error Output Voltage Range Output Resistance OUTPUT BUFFER Gain Gain Error Output Voltage Range Input Bias Current Output Resistance DYNAMIC RESPONSE System Bandwidth Slew Rate NOISE 0.1 Hz to 10 Hz Spectral Density, 1 kHz (RTI) POWER SUPPLY Operating Range Quiescent Current vs. Temperature PSRR TEMPERATURE RANGE For Specified Performance
1 2
7 -0.3 0.02 97 +0.3 4.8 103 -0.3 0.02 97
7 +0.3 4.8 103 -0.3 0.02 97
7 +0.3 4.8 103
100 2
100 2
100 2
V/V % V k V/V % V nA kHz V/s V p-p nV/Hz
0.02 VOUT 4.8 V dc
-0.3 0.02 40 2
+0.3 4.8
-0.3 0.02 40 2 40 60 0.33 10 300
+0.3 4.8
-0.3 0.02 40 2 40 60 0.33 10 300
+0.3 4.8
VIN = 0.01 V p-p, VOUT = 0.14 V p-p VIN = 0.28 V, VOUT = 4 V step
40
60 0.33 10 300
3.5 VO = 0.1 V dc VS = 3.5 V to 12 V 75 -40 0.25 83
12 1.0
3.5 0.25 75 83
12 1.0
3.5 0.25 75 83
12 1.0
V mA dB
+125
-40
+125
-40
+150
C
Source imbalance <2 . The AD8203 preamplifier exceeds 80 dB CMRR at 10 kHz. However, since the signal is available only by way of a 100 k resistor, even the small amount of pin-to-pin capacitance between Pin 1, Pin 8 and Pin 3, Pin 4 may couple an input common-mode signal larger than the greatly attenuated preamplifier output. The effect of pinto-pin coupling may be neglected in all applications by using filter capacitors at Node 3.
Rev. B | Page 3 of 20
AD8203 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Voltage Transient Input Voltage (400 ms) Continuous Input Voltage (Common Mode) Reversed Supply Voltage Protection Operating Temperature Range Die SOIC MSOP Storage Temperature Output Short-Circuit Duration Lead Temperature Range (Soldering 10 sec) Rating 12.5 V 44 V 35 V 0.3 V -40C to +150C -40C to +125C -40C to +125C -65C to +150C Indefinite 300C
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. B | Page 4 of 20
AD8203 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
-IN 1 GND 2 A1 3
8
+IN +VS
AD8203
7
NC = NO CONNECT
Figure 4. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. 1 2 3 4 5 6 7 8 Mnemonic -IN GND A1 A2 OUT NC +VS +IN X -409.0 -244.6 +229.4 +410.0 +410.0 NA +121.0 -409.0 Y -205.2 -413.0 -413.0 -308.6 +272.4 NA +417.0 +205.2
05013-004
6 NC TOP VIEW A2 4 (Not to Scale) 5 OUT
1036m +VS
OUT +IN
1048m
-IN A2
GND
A1
Figure 5. Metallization Photograph
Rev. B | Page 5 of 20
05013-005
AD8203 TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25C, VS = 5 V, VCM = 0 V, RL = 10 k, unless otherwise noted.
90 80 -5 70 60 0
COMMON-MODE VOLTAGE (V)
-55C -40C
-10
PSRR (dB)
50 40 30 20 10 0 10 100 1k FREQUENCY (Hz) 10k
05013-006
-15
+25C
-20
+125C
-25 +150C -30 3 4 5 6 7 8 9 POWER SUPPLY (V) 10 11
05013-009
100k
12
Figure 6. Power Supply Rejection Ratio vs. Frequency for Common-Mode Range -6 V to +30 V
25
Figure 9. Negative Common-Mode Voltage vs. Voltage Supply
40
35 20
COMMON-MODE VOLTAGE (V)
+25C
30 -55C 25 +150C 20
OUTPUT (dB)
15
10
5
05013-007
15 +125C 10 3 -40C 4 5 6 7 8 9 POWER SUPPLY (V) 10 11
05013-010
0 100
1k
10k FREQUENCY (Hz)
100k
1M
12
Figure 7. Bandwidth
100 95 90 85
Figure 10. Positive Common-Mode Voltage vs. Voltage Supply
5.0
4.0
OUTPUT VOLTAGE (dB)
05013-008
CMRR (dB)
80 75 70 65 60 55 50 10 100 1k FREQUENCY (Hz) 10k
3.0
2.0
1.0
05013-011
100k
0 10
100 1k LOAD RESISTANCE ()
10k
Figure 8. Common-Mode Rejection Ratio vs. Frequency for Common-Mode Range -6 V to +30 V
Figure 11. Output Swing vs. Load Resistance
Rev. B | Page 6 of 20
AD8203
0
40 -6V TO +30V COMMON MODE TEMPERATURE = 25C 35
-10
OUTPUT MINUS SUPPLY (mV)
-20
NO LOAD
30 25
-30
HITS
-40 10k LOAD -50
05013-012
20 15 10
05013-051
-60 -70 3 4 5 6 7 8 9 10 SUPPLY VOLTAGE (V) 11 12
5 0
13
Figure 12. Swing Minus Supply vs. Supply Voltage
7
OUTPUT
6
5
HITS
4
3
4
INPUT
2 1
05013-013
-80 -72 -64 -56 -48 -40 -32 -24 -16 -8 0 8 16 24 32 40 48 56 64 72 80
CMRR (V/V)
Figure 15. CMRR Distribution, Temperature = 25C
VSUPPLY = 5V TEMPERATURE RANGE = +25C TO -40C
3 CH3 100mV CH4 1.0V M 20s 2.5MS/s 400NS/PT A CH3 260mV
Figure 13. Pulse Response
8 7
-40C +25C 400
1000 800 600
6 5
HITS
VOS (V)
200 0 -200 +85C -400 -600
05013-052
4 3
+125C
2
05013-026
-800 -1000 -10 -5 0 5 10 15 20 25 COMMON-MODE VOLTAGE (V) 30
1 0
35
Figure 14. VOS vs. Common-Mode Voltage
Rev. B | Page 7 of 20
-30 -28 -26 -24 -22 -20 -18 -16 -14 -12 -10 -8 -6 -4 -2 0 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30
-30 -28 -26 -24 -22 -20 -18 -16 -14 -12 -10 -8 -6 -4 -2 0 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30
0
VOS DRIFT (V/C)
Figure 16. Offset Drift Distribution, MSOP, Temperature Range = +25C to -40C
VSUPPLY = 5V TEMPERATURE RANGE = 25C TO 85C
VOS DRIFT (V/C)
Figure 17. Offset Drift Distribution, MSOP, Temperature Range = 25C to 85C
05013-025
AD8203
9 8 7 6
HITS
8
VSUPPLY = 5V TEMPERATURE RANGE = 25C TO 125C
PACKAGE = MSOP @ -40C 7 6 5
HITS
05013-027
5 4 3 2 1 0
4 3 2
05013-030
1 0
-30 -28 -26 -24 -22 -20 -18 -16 -14 -12 -10 -8 -6 -4 -2 0 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30
VOS DRIFT (V/C)
Figure 18. VOS Distribution, MSOP, Temperature Range = 25C to 125C
10 PACKAGE = MSOP @ 25C 9 8
8 TEMPERATURE = 25C 7 6
7 6
HITS HITS 5 4 3 2
05013-031
5 4 3 2 1
-2200 -2000 -1800 -1600 -1400 -1200 -1000 -800 -600 -400 -200 0 200 400 600 800 1000 1200 1400 1600 1800 2000 2200
05013-028
1 0
0.02
0.06
0.10
0.14
0.18
0.22
0.26
0.04
0.08
0.12
0.16
0.20
0.24
VOS (V)
ERROR (%)
Figure 19. VOS Distribution, MSOP, Temperature = 25C
14 PACKAGE = MSOP @ 125C 12
6 5 7
Figure 22. MSOP Gain Accuracy, Temperature = 25C
TEMPERATURE = 125C
10
8
HITS
4
6
HITS
3 2
4
05013-029
0.02
0.06
0.10
0.14
0.18
0.22
0.26
0.04
0.08
0.12
0.16
0.20
0.24
-2200 -2000 -1800 -1600 -1400 -1200 -1000 -800 -600 -400 -200 0 200 400 600 800 1000 1200 1400 1600 1800 2000 2200
VOS (V)
ERROR (%)
Figure 20. VOS Distribution, MSOP, Temperature = 125C
Figure 23. MSOP Gain Accuracy, Temperature = 125C
Rev. B | Page 8 of 20
0.28
0.30
0
0
0
05013-032
2
1
0.28
0.30
0
0
-2200 -2000 -1800 -1600 -1400 -1200 -1000 -800 -600 -400 -200 0 200 400 600 800 1000 1200 1400 1600 1800 2000 2200
VOS (V)
Figure 21. VOS Distribution, MSOP, Temperature = -40C
AD8203
7 TEMPERATURE = -40C 6
18 16 14 PACKAGE = MSOP VSUPPLY = 5V TEMPERATURE RANGE = 25C TO 125C
5
12
HITS
4
HITS
10 8 6
3
2
4
05013-033
2
0.04
0.08
0.12
0.16
0.20
0.24
0.02
0.06
0.10
0.14
0.18
0.22
0.26
0.28
0.30
0
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
2
3
4
5
6
7
8
ERROR (%)
GAIN DRIFT (ppm/C)
Figure 24. MSOP Gain Accuracy, Temperature = -40C
Figure 27. Gain Drift Distribution, MSOP, Temperature Range = 25C to 125C
14 PACKAGE = SOIC @ 25C 12
12
PACKAGE = MSOP VSUPPLY = 5V TEMPERATURE RANGE = +25C TO -40C
10
10
8
8 HITS
6
HITS
6
4
4
2
05013-036
GAIN DRIFT (ppm/C)
Figure 25. Gain Drift Distribution, Temperature Range = +25C to -40C
12 PACKAGE = MSOP VSUPPLY = 5V TEMPERATURE RANGE = 10 25C TO 85C
9 PACKAGE = SOIC @ 125C 8 7
8
6 HITS 5 4 3 2
HITS
6
4
2
05013-037
-2000 -1800 -1600 -1400 -1200 -1000 -800 -600 -400 -200 0 200 400 600 800 1000 1200 1400 1600 1800 2000 VOS (V)
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
2
3
4
5
6
7
8
9
0
0
Figure 28. VOS Distribution, SOIC, Temperature = 25C
1 -2000 -1800 -1600 -1400 -1200 -1000 -800 -600 -400 -200 0 200 400 600 800 1000 1200 1400 1600 1800 2000 VOS (V) 0
-9
-8
-7
-6
-5
-4
-3
-2
-1
0
1
2
3
4
5
6
7
8
GAIN DRIFT (ppm/C)
Figure 26. Gain Drift Distribution, MSOP, Temperature Range = 25C to 85C
9
0
Figure 29. VOS Distribution, SOIC, Temperature = 125C
Rev. B | Page 9 of 20
05013-040
05013-039
2
9
0
0
05013-038
1
AD8203
14 PACKAGE = SOIC @ -40C 12 5 6 PACKAGE = SOIC VSUPPLY = 5V TEMPERATURE RANGE = 25C TO 125C
10 4 8 HITS HITS 6 2 4 1
05013-041 05013-044
3
2 0
-15.0 -13.5 -12.0 -10.5 -9.0 -7.5 -6.0 -4.5 -3.0 -1.5 0 1.5 3.0 4.5 6.0 7.5 9.0 10.5 VOS DRIFT (mV/C)
-2000 -1800 -1600 -1400 -1200 -1000 -800 -600 -400 -200 0 200 400 600 800 1000 1200 1400 1600 1800 2000
VOS (V)
Figure 30. VOS Distribution, SOIC, Temperature = -40C
6
9
Figure 33. Offset Drift Distribution, SOIC, Temperature Range = +25C to 125C
TEMPERATURE = 25C 8 7
5
PACKAGE = SOIC VSUPPLY = 5V TEMPERATURE RANGE = +25C TO -40C
4 HITS HITS
6 5 4 3 2
3
2
1
05013-042
12.0 13.5 15.0
05013-046 05013-045
0
1
-15.0
-13.5 -12.0 -10.5 -9.0 -7.5 -6.0 -4.5 -3.0 -1.5 0 1.5 3.0 4.5 6.0 7.5 9.0 10.5
VOS DRIFT (V/C)
Figure 31. Offset Drift Distribution, SOIC, Temperature Range = +25C to -40C
6 PACKAGE = SOIC VSUPPLY = 5V TEMPERATURE RANGE = 25C TO 85C
9
8 7
5
4 HITS HITS
6 5 4 3 2
3
2
1
05013-043
1
-15.0
-13.5 -12.0 -10.5 -9.0 -7.5 -6.0 -4.5 -3.0 -1.5 0 1.5 3.0 4.5 6.0 7.5 9.0 10.5
12.0 13.5 15.0
VOS DRIFT (V/C)
Figure 32. Offset Drift Distribution, SOIC, Temperature Range = 25C to 85C
Rev. B | Page 10 of 20
0 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11 0.12 0.13 0.14 0.15 0.16 0.17 0.18 0.19 0.20 0.21 0.22 0.23 0.24 0.25 0.26 0.27 0.28 0.29 0.30
ERROR (%)
0
0
0 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11 0.12 0.13 0.14 0.15 0.16 0.17 0.18 0.19 0.20 0.21 0.22 0.23 0.24 0.25 0.26 0.27 0.28 0.29 0.30
ERROR (%)
12.0 13.5 15.0
0
0
Figure 34. Gain Accuracy, SOIC, Temperature = 25C
TEMPERATURE = 125C
Figure 35. Gain Accuracy, SOIC, Temperature = 125C
AD8203
12 TEMPERATURE = -40C 10 10 PACKAGE = SOIC 9 VSUPPLY = 5V TEMPERATURE RANGE = 8 25C TO 85C 7 6
8
HITS
6
HITS
05013-047
5 4
4
3 2 1
05013-049
2
0 0.01 0.02 0.03 0.04 0.05 0.06 0.07 0.08 0.09 0.10 0.11 0.12 0.13 0.14 0.15 0.16 0.17 0.18 0.19 0.20 0.21 0.22 0.23 0.24 0.25 0.26 0.27 0.28 0.29 0.30
0
2
4
6
-8
-6
-4
-2
8
10
12
14
-16
-14
-12
ERROR (%)
-10
GAIN DRIFT (ppm/C)
Figure 36. Gain Accuracy, SOIC, Temperature = -40C
10 9 8 7 6 PACKAGE = SOIC VSUPPLY = 5V TEMPERATURE RANGE = +25C to -40C
Figure 38. Gain Drift Distribution, SOIC, Temperature Range = 25C to 85C
10 9 8 7 6 HITS 5 4 3 2
05013-048
PACKAGE = SOIC VSUPPLY = 5V TEMPERATURE RANGE = 25C TO 125C
HITS
5 4 3 2 1
16
05013-050
0
0
1 -12 -11 -10 -9 -8 -7 -6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6 7 8 9 10 11 12 GAIN DRIFT (ppm/C) 0
0
2
4
6
-8
-6
-4
-2
8
10
12
14
-16
-14
-12
-10
GAIN DRIFT (ppm/C)
Figure 37. Gain Drift Distribution, SOIC, Temperature Range = +25C to -40C
16
0
Figure 39. Gain Drift Distribution, SOIC, Temperature Range = 25C to 125C
Rev. B | Page 11 of 20
AD8203 THEORY OF OPERATION
The AD8203 consists of a preamp and buffer, arranged as shown in Figure 40. Like-named resistors have equal values. The preamp incorporates a dynamic bridge (subtractor) circuit. Identical networks (within the shaded areas) consisting of RA, RB, RC, and RG, attenuate input signals applied to Pin 1 and Pin 8. Note that when equal amplitude signals are asserted at Input 1 and Input 8, and the output of A1 is equal to the common potential (that is, 0), the two attenuators form a balanced-bridge network. When the bridge is balanced, the differential input voltage at A1, and thus its output, is 0.
B
A3 amplifier detects the common-mode signal applied to A1 and adjusts the voltage on the matched RCM resistors to reduce the common-mode voltage range at the A1 inputs. By adjusting the common voltage of these resistors, the common-mode input range is extended while, at the same time, the normal mode signal attenuation is reduced, leading to better performance referred to input. The output of the dynamic bridge taken from A1 is connected to Pin 3 by way of a 100 k series resistor, provided for lowpass filtering and gain adjustment. The resistors in the input networks of the preamp and the buffer feedback resistors are ratio-trimmed for high accuracy. The output of the preamp drives a gain-of-2 buffer amplifier, A2, implemented with carefully matched feedback resistors RF. The 2-stage system architecture of the AD8203 enables the user to incorporate a low-pass filter prior to the output buffer. By separating the gain into two stages, a full-scale, rail-to-rail signal from the preamp can be filtered at Pin 3, and a half-scale signal, resulting from filtering, can be restored to full scale by the output buffer amp. The source resistance seen by the inverting input of A2 is approximately 100 k to minimize the effects of the input bias current of A2. However, this current is quite small, and errors resulting from applications that mismatch the resistance are correspondingly small. The A2 input bias current has a typical value of 40 nA, however, this can increase under certain conditions. For example, if the input signal to the A2 amplifier is VCC/2, the output attempts to go to VCC due to the gain of 2. However, the output saturates because the maximum specified voltage for correct operation is 200 mV below VCC. Under these conditions the total input bias current increases (see Figure 41 for more information).
-140
Any common-mode voltage applied to both inputs keeps the bridge balanced and the A1 output at 0. Because the resistor networks are carefully matched, the common-mode signal rejection approaches this ideal state. However, if the signals applied to the inputs differ, the result is a difference at the input to A1. A1 responds by adjusting its output to drive RB, by way of RG, to adjust the voltage at its inverting input until it matches the voltage at its noninverting input.
B
By attenuating voltages at Pin 1 and Pin 8, the amplifier inputs are held within the power supply range, even if Pin 1 and Pin 8 input levels exceed the supply or fall below common (ground). The input network also attenuates normal (differential) mode voltages. RC and RG form an attenuator that scales A1 feedback, forcing large output signals to balance relatively small differential inputs. The resistor ratios establish the preamp gain at 7. Because the differential input signal is attenuated and then amplified to yield an overall gain of 7, Amplifier A1 operates at a higher noise gain, multiplying deficiencies such as input offset voltage and noise with respect to Pin 1 and Pin 8.
+IN
8
-IN
1
RA
RA 100k (TRIMMED) RCM RCM A3 RF RG A2 RF
5
-120
A2 INPUT BIAS CURRENT (nA)
A1
3
4
-100
RB RG RC
RB RC
-80
AD8203
05013-014
-60 -40
2
COM
Figure 40. Simplified Schematic
To minimize these errors while extending the common-mode range, a dedicated feedback loop is used to reduce the range of common-mode voltage applied to A1 for a given overall range at the inputs. By offsetting the range of voltage applied to the compensator, the input common-mode range is also offset to include voltages more negative than the power supply. The
0
0
0.5 1.0 1.5 2.0 DIFFERENTIAL MODE VOLTAGE (V)
2.5
Figure 41. A2 Input Bias Current vs. Input Voltage and Temperature. The Shaded Area Is the Bias Current from -40C to +125C.
An increase in the A2 bias current, in addition to the output saturation voltage of A1, directly affects the output voltage of
Rev. B | Page 12 of 20
05013-035
-20
AD8203
the AD8203 system (Pin 3 and Pin 4 shorted). An example of how to calculate the correct output voltage swing of the AD8203, by taking all variables into account, follows: * * Amplifier A1 output saturation potential can go as low as 20 mV at its output. A2 typical input bias current of 40 nA multiplied by the 100 k preamplifier output resistor produces 40 nA x 100 k = 4 mV at the A2 input * Total voltage at the A2 input equals the output saturation voltage of A1 combined with the voltage error generated by the input bias current 20 mV + 4 mV = 24 mV * The total error at the input of A2, 24 mV, multiplied by the buffer gain generates a resulting error of 48 mV at the output of the buffer. This is the AD8203 system output low saturation potential. The high output voltage range of the AD8203 is specified as 4.8 V. Therefore, assuming a typical A2 input bias current, the output voltage range for the AD8203 is 48 mV to 4.8 V.
*
For an example of the effect of changes in A2 input bias current vs. applied input potentials, see Figure 41. The change in bias current causes a change in error voltage at the input of the buffer amplifier. This results in a change in overall error potential at the output of the buffer amplifier.
Rev. B | Page 13 of 20
AD8203 APPLICATIONS
The AD8203 difference amplifier is intended for applications that require extracting a small differential signal in the presence of large common-mode voltages. The input resistance is nominally 320 k, and the device can tolerate common-mode voltages higher than the supply voltage and lower than ground. The open collector output stage sources current to within 20 mV of ground and to within 200 mV of VS.
VCM +VS OUT
+IN +VS NC OUT
VDIFF 2 VDIFF 2
10k
10k GAIN =
14REXT REXT + 100k GAIN 14 - GAIN
AD8203
100k
REXT = 100k
-IN
GND
A1
A2
CURRENT SENSING
Basic automotive applications making use of the large commonmode range are shown in Figure 2 and Figure 3. The capability of the device to operate as an amplifier in primary battery supply circuits is shown in Figure 2. Figure 3 illustrates the ability of the device to withstand voltages below system ground.
NC = NO CONNECT
05013-016
High Line, High Current Sensing
REXT
Figure 43. Adjusting for Gains < 14
Low Current Sensing
The AD8203 is also used in low current sensing applications, such as the 4 to 20 mA current loop shown in Figure 42. In such applications, the relatively large shunt resistor can degrade the common-mode rejection. Adding a resistor of equal value on the low impedance side of the input corrects this error.
10 1%
+IN
The overall bandwidth is unaffected by changes in gain by using this method, although there may be a small offset voltage due to the imbalance in source resistances at the input to the buffer. This can often be ignored, but if desired, it can be nulled by inserting a resistor equal to 100 k minus the parallel sum of REXT and 100 k, in series with Pin 4. For example, with REXT = 100 k (yielding a composite gain of x7), the optional offset nulling resistor is 50 k.
Gains Greater Than 14
5V OUTPUT
+VS NC OUT
+
10 1%
AD8203
-IN GND A1 A2
Connecting a resistor from the output of the buffer amplifier to its noninverting input, as shown in Figure 44, increases the gain. The gain is now multiplied by the factor REXT/(REXT - 100 k); for example, the gain is doubled for REXT = 200 k. Overall gains as high as 50 are achievable this way. Note that the accuracy of the gain becomes critically dependent on the resistor value at high gains. Also, the effective input offset voltage at Pin 1 and Pin 8 (about six times the actual offset of A1) limits the part's use in high gain, dc-coupled applications.
+VS OUT
+IN +VS NC OUT
NC = NO CONNECT
Figure 42. 4 to 20 mA Current Loop Receiver
VDIFF 2 VDIFF 2
05013-015
10k
10k GAIN = REXT
14REXT REXT - 100k GAIN GAIN - 14
GAIN ADJUSTMENT
The default gain of the preamplifier and buffer are x7 and x2, respectively, resulting in a composite gain of x14. With the addition of external resistor(s) or trimmer(s), the gain can be lowered, raised, or finely calibrated.
VCM
AD8203
100k
REXT = 100k
-IN
GND
A1
A2
NC = NO CONNECT
Gains Less Than 14
Since the preamplifier has an output resistance of 100 k, an external resistor connected from Pin 3 and Pin 4 to GND decreases the gain by a factor REXT/(100 k + REXT), as shown in Figure 43.
Figure 44. Adjusting for Gains > 14
Rev. B | Page 14 of 20
05013-017
AD8203
GAIN TRIM
Figure 45 shows a method for incremental gain trimming by using a trim potentiometer and external resistor REXT. The following approximation is useful for small gain ranges: G (10 M/REXT)% Thus, the adjustment range is 2% for REXT = 5 M; 10% for REXT = 1 M, and so on.
5V OUT
+IN +VS NC OUT
Low-pass filters can be implemented in several ways by using the features provided by the AD8203. In the simplest case, a single-pole filter (20 dB/decade) is formed when the output of A1 is connected to the input of A2 via the internal 100 k resistor by strapping Pin 3, Pin 4, and a capacitor added from this node to ground, as shown in Figure 46. If a resistor is added across the capacitor to lower the gain, the corner frequency increases; it should be calculated using the parallel sum of the resistor and 100 k.
5V OUTPUT
+IN +VS NC OUT
VDIFF 2
VDIFF 2
AD8203
VCM VDIFF 2
-IN GND A1 A2
AD8203
VCM VDIFF 2
-IN GND A1 A2
fC =
1 2C105
C IN FARADS
REXT
GAIN TRIM 20k MIN
C
05013-018
NC = NO CONNECT
NC = NO CONNECT
Figure 45. Incremental Gain Trim
Figure 46. Single-Pole, Low-Pass Filter Using the Internal 100 k Resistor
Internal Signal Overload Considerations
When configuring gain for values other than 14, the maximum input voltage with respect to the supply voltage and ground must be considered, since either the preamplifier or the output buffer reaches its full-scale output (approximately VS - 0.2 V) with large differential input voltages. The input of the AD8203 is limited to (VS - 0.2)/7 for overall gains 7, since the preamplifier, with its fixed gain of x7, reaches its full-scale output before the output buffer. For gains greater than 7, the swing at the buffer output reaches its full scale first and limits the AD8203 input to (VS - 0.2)/G, where G is the overall gain.
If the gain is raised using a resistor, as shown in Figure 44, the corner frequency is lowered by the same factor as the gain is raised. Thus, using a resistor of 200 k (for which the gain would be doubled), the corner frequency is now 0.796 Hz F (0.039 F for a 20 Hz corner frequency).
5V OUT
+IN +VS NC OUT
VDIFF 2
AD8203
VCM VDIFF 2
-IN GND A1 A2
C
LOW-PASS FILTERING
In many transducer applications, it is necessary to filter the signal to remove spurious high frequency components, including noise, or to extract the mean value of a fluctuating signal with a peak-to-average ratio (PAR) greater than unity. For example, a full-wave rectified sinusoid has a PAR of 1.57, a raised cosine has a PAR of 2, and a half-wave sinusoid has a PAR of 3.14. Signals having large spikes can have PARs of 10 or more. When implementing a filter, the PAR should be considered so that the output of the AD8203 preamplifier (A1) does not clip before A2, since this nonlinearity would be averaged and appear as an error at the output. To avoid this error, both amplifiers should be made to clip at the same time. This condition is achieved when the PAR is no greater than the gain of the second amplifier (2 for the default configuration). For example, if a PAR of 5 is expected, the gain of A2 should be increased to 5.
255k C NC = NO CONNECT
005013-020
fC(Hz) = 1/C(F)
Figure 47. 2-Pole, Low-Pass Filter
A 2-pole filter (with a roll-off of 40 dB/decade) can be implemented using the connections shown in Figure 47. This is a Sallen-Key form based on a x2 amplifier. It is useful to remember that a 2-pole filter with a corner frequency f2 and a 1-pole filter with a corner at f1 have the same attenuation at the frequency (f22/f1). The attenuation at that frequency is 40 log (f2/f1), which is illustrated in Figure 48. Using the standard resistor value shown and equal capacitors (see Figure 47), the corner frequency is conveniently scaled at 1 Hz F (0.05 F for a 20 Hz corner). A maximally flat response occurs when the resistor is lowered to 196 k and the scaling is then 1.145 Hz F. The output offset is raised by approximately 5 mV (equivalent to 250 V at the input pins).
Rev. B | Page 15 of 20
05013-019
AD8203
FREQUENCY
ATTENUATION
40dB/DECADE 20dB/DECADE
by a 1-pole low-pass filter, shown in Figure 49, set with a corner frequency of 3.6 Hz, which provides about 30 dB of attenuation at 100 Hz. A higher rate of attenuation can be obtained using a 2-pole filter with fC = 20 Hz, as shown in Figure 50. Although this circuit uses two separate capacitors, the total capacitance is less than half that needed for the 1-pole filter.
INDUCTIVE 5V LOAD OUTPUT
+IN +VS NC OUT
40log (f2/f1)
CLAMP DIODE
05013-021
A 1-POLE FILTER, CORNER f1, AND A 2-POLE FILTER, CORNER f2, HAVE THE SAME ATTENUATION -40log (f2/f1) AT FREQUENCY f22/f1 f1 f2 f22/f1
BATTERY
14V 4-TERM SHUNT
301k
AD8203
-IN GND A1 A2
C 50k
Figure 48. Comparative Responses of 1-Pole and 2-Pole Low-Pass Filters
POWER DEVICE 93k C NC = NO CONNECT COMMON fC(Hz) = 1/C(F) (0.05F FOR fC = 20Hz)
05013-023
HIGH LINE CURRENT SENSING WITH LPF AND GAIN ADJUSTMENT
Figure 49 is another refinement of Figure 2, including gain adjustment and low-pass filtering.
CLAMP DIODE INDUCTIVE 5V LOAD OUT 4V/AMP
NC OUT
Figure 50. 2-Pole Low-Pass Filter
DRIVING CHARGE REDISTRIBUTION ADCS
When driving CMOS ADCs, such as those embedded in popular microcontrollers, the charge injection (Q) can cause a significant deflection in the output voltage of the AD8203. Though generally of short duration, this deflection may persist until after the sample period of the ADC has expired due to the relatively high open-loop output impedance (21 k) of the AD8203. Including an R-C network in the output can significantly reduce the effect. The capacitor helps to absorb the transient charge, effectively lowering the high frequency output impedance of the AD8203. For these applications, the output signal should be taken from the midpoint of the RLAG to CLAG combination, as shown in Figure 51. Since the perturbations from the analog-to-digital converter are small, the output impedance of the AD8203 appears to be low. The transient response, therefore, has a time constant governed by the product of the two LAG components, CLAG x RLAG. For the values shown in Figure 51, this time constant is programmed at approximately 10 s. Therefore, if samples are taken at several tens of microseconds or more, there is negligible charge stack-up.
5V
4 7
+IN
+VS
BATTERY
14V 4-TERM SHUNT
133k
AD8203
20k
-IN GND A1 A2
POWER DEVICE
VOS/IB NULL C
NC = NO CONNECT
COMMON
5% CALIBRATION RANGE fC(Hz) = 0.767Hz/C(F) (0.22F FOR fC = 3.6Hz)
Figure 49. High Line Current Sensor Interface; Gain = x40, Single-Pole Low-Pass Filter
A power device that is either on or off controls the current in the load. The average current is proportional to the duty cycle of the input pulse and is sensed by a small value resistor. The average differential voltage across the shunt is typically 100 mV, although its peak value is higher by an amount that depends on the inductance of the load and the control frequency. The common-mode voltage, conversely, extends from roughly 1 V above ground for the on condition to about 1.5 V above the battery voltage for the off condition. The conduction of the clamping diode regulates the common-mode potential applied to the device. For example, a battery spike of 20 V may result in an applied common-mode potential of 21.5 V to the input of the devices. To produce a full-scale output of 4 V, a gain x40 is used, adjustable by 5% to absorb the tolerance in the shunt. There is sufficient headroom to allow 10% overrange (to 4.4 V). The roughly triangular voltage across the sense resistor is averaged
05013-022
+IN
AD8203
A2
5
RLAG 1k CLAG 0.01F
MICROPROCESSOR A/D
-IN
10k
10k
2
05013-024
Figure 51. Recommended Circuit for Driving CMOS A/D
Rev. B | Page 16 of 20
AD8203 OUTLINE DIMENSIONS
3.20 3.00 2.80
5.00 (0.1968) 4.80 (0.1890)
8 5 4
3.20 3.00 2.80
8
5
1
5.15 4.90 4.65
4.00 (0.1574) 3.80 (0.1497) 1
6.20 (0.2440) 5.80 (0.2284)
4
1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040)
1.10 MAX 8 0 0.80 0.60 0.40
PIN 1 0.65 BSC 0.95 0.85 0.75 0.15 0.00 0.38 0.22 SEATING PLANE
1.75 (0.0688) 1.35 (0.0532)
0.50 (0.0196) x 45 0.25 (0.0099)
0.51 (0.0201) COPLANARITY SEATING 0.31 (0.0122) 0.10 PLANE
8 0.25 (0.0098) 0 1.27 (0.0500) 0.40 (0.0157) 0.17 (0.0067)
0.23 0.08
COPLANARITY 0.10
COMPLIANT TO JEDEC STANDARDS MS-012-AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MO-187-AA
Figure 52. 8-Lead Mini Small Outline Package [MSOP] (RM-8) Dimensions shown in millimeters
Figure 53. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model AD8203YRMZ 1 AD8203YRMZ-RL1 AD8203YRMZ-R71 AD8203YRZ1 AD8203YRZ-RL1 AD8203YRZ-R71 AD8203YCSURF
1
Temperature Package -40C to +125C -40C to +125C -40C to +125C -40C to +125C -40C to +125C -40C to +125C
Package Description 8-Lead Mini Small Outline Package [MSOP] 8-Lead Mini Small Outline Package [MSOP] 8-Lead Mini Small Outline Package [MSOP] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] Die
Package Outline RM-8 RM-8 RM-8 R-8 R-8 R-8
Branding JXA JXA JXA
Z = Pb-free part.
Rev. B | Page 17 of 20
AD8203 NOTES
Rev. B | Page 18 of 20
AD8203 NOTES
Rev. B | Page 19 of 20
AD8203 NOTES
(c) 2005 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05013-0-10/05(B)
Rev. B | Page 20 of 20

▲Up To Search▲

Price & Availability of AD8203

	To Download AD8203 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .